Transmission line, resonator, filter, duplexer, and communication apparatus

ABSTRACT

A transmission line, a resonator, a filter, a duplexer, and a communication apparatus efficiently minimize power losses due to edge effects, thereby having superior loss-reduction characteristics. A continuous line and a plurality of thin lines each having a predetermined length and branching from both sides of the continuous line are formed on a dielectric substrate. With this structure, edges of the individual thin lines substantially do not exist, so that losses due to edge effects can be efficiently minimized.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a divisional of application Ser. No.09/556,120, filed Apr. 19, 2000, in the name of Seiji HIDAKA, MichiakiOTA and Shin ABE, now U.S. Pat. No. 6,633,207.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a transmission line, a resonator, afilter, a duplexer, and a communication apparatus used for radiocommunication and for transmitting and receiving electromagnetic wavesin, for example, microwave bands and millimeter-wave bands.

2. Description of the Related Art

Ordinary RF circuits employ planar circuits that use transmission lines,such as microstrip lines, which can be easily produced and that aresuitable for being miniaturized and made thin.

In the microstrip line, however, current concentration due to surfaceeffects occurs on a conductor surface. Particularly, it is apparent atthe edges, causing a power loss in a narrow region in a range of severalmicrometers (μm) to several tens of micrometers (μm) around the edges,accounting for 50% of the entire power loss. This phenomenon, called anedge effect, is attributed to the cross-sectional shape of the conductor(electrode). In planar circuits in which electrodes, such as microstriplines, are formed on a substrate, edges always exist. Therefore, theproblem of power loss due to the edge effect always occurs and is knownto be unavoidable.

In this connection, RF transmission lines for aiming to reduce thecurrent concentration at the conductor edges were suggested as disclosedin (1) Japanese Unexamined Patent Application Publication No. 8-321706and (2) Japanese Unexamined Patent Application Publication No. 10-13112.

In each of the above publications, a plurality of linear conductors isformed at a constant pitch, parallel to a signal-propagation direction.It can be said that, in the above-described conventional transmissionlines, the conductor is divided parallel to the signal-propagationdirection to reduce the current concentration at the edges. However, toform the conductors with the correct line width in these structuresrequires very severe manufacturing accuracy, on the same order ofmagnitude as the skin depth. In addition, the conductor Q value isimproved only within a small range of 10 to 20% of the conventional Qvalue. Further, depending on the dividing method, there are cases wherethe conductor Q value decreases, so that it is actually lower than the Qvalue of a single-line conductor.

Thus, in the structure in which the direction of a current path is thesame as the signal-propagation direction, even when the linewidth isdivided to be as thin as possible, the left and right edges still exist.Therefore, the structures are not effective enough as a fundamentalsolution to the edge effect problem.

SUMMARY OF THE INVENTION

In view of the above, an object of the present invention is to provide atransmission line, a resonator, a filter, a duplexer, and acommunication apparatus that efficiently minimize power losses due toedge effects, thereby having superior loss-reduction characteristics.

In order to achieve the above object, a transmission line of anembodiment of the present invention is configured of at least onecontinuous line and a plurality of thin lines each branching from thecontinuous line and having a predetermined length.

According to this structure, other thin lines having the same shape maybe arranged adjacent to the one thin line. In this case, since physicaledges exist when microscopically viewed, a weak edge effect occurs atthe edge of each of the thin lines. However, when the plurality of thelines are macroscopically viewed as a whole, the edge on the left sideof one of the connected thin lines exists adjacent to, for example, theright edge of another one of the connected thin lines. Therefore,substantial edges in the line width direction do not exist; that is, theedge is not noticeable. This allows the current concentration at theedges of the lines to be efficiently reduced, thereby minimizing theentire power loss.

The thin lines do not deteriorate the transmission characteristics ofthe transmission line since the successive thin lines together functionas a single high-frequency transmission line. Because respective ends ofthe thin lines are connected to a common continuous line, currenttransmitted in the continuous line flows into the respectivetransmission lines. Therefore, magnetic fields are induced around therespective thin lines. Due to the coupling between the magnetic fields,the thin lines are electromagnetically coupled with each other. As aresult, a high frequency signal can be propagated via the successivethin lines along the extension direction of the continuous line.

Also, in the transmission line of the present invention, the branchingdirection of each of the thin lines may be slanted with respect to thecontinuous line. In this case, the direction in which the thin linesextend has a component extending in the signal-propagation direction forall of the lines, thereby allowing the edge effect to be efficientlyminimized.

Also, in the transmission line of the present invention, theaforementioned individual thin lines may be connected, and theaforementioned continuous line connects identical portions of theaforementioned lines. For example, each of the thin lines is arranged tohave substantially an integer multiple length of half the wavelengthcorresponding to the transmission frequency, and central portions of theindividual thin lines are connected. With this arrangement, both ends ofeach of the thin lines become open ends, and portions that representnodes in the voltage amplitudes are connected via the continuous line.Alternatively, by connecting both ends of each of the thin lines via thecontinuous line, both ends of each of the thin lines becomeshort-circuited ends, and portions that represent antinodes in thecurrent amplitudes are connected.

According to these structures, electromagnetic-field distributions(voltage and current distributions) on the individual thin lines areforced by the continuous line to be uniform. This increases theefficiency of the reduction of the edge effect due to the closearrangement of the individual thin lines.

Also, in the transmission line of the present invention, the thin linesmay be curved lines, and a controlled capacitive coupling or mutualdielectric coupling between each pair of thin lines may be arranged.

Also, in the transmission line of the present invention, a line width ofeach of the thin lines may be not more than the skin depth of aconductor of each of the lines. By this structure, currents that flow tomaintain magnetic fields that pass through gaps between left sides andright sides of the individual lines are spaced apart by suitabledistances so as to cause interference at the left sides and the rightsides thereof. This minimizes reactive currents deviating in phase,thereby allowing the power loss to be significantly reduced.

Also, in the transmission line of the present invention, each of thethin lines may be a thin-film multilayered electrode, having overlaidthin-film dielectric layers and thin-film conductor layers. By this, theskin effect in the direction from the substrate surface to the outsideof the electrode can be reduced. This allows a further reduction in thepower loss to be obtained.

Also, in the transmission line of the present invention, a dielectricmaterial may be filled in each gap between the adjacent thin lines. Bythis, short-circuiting between lines is prevented, including when thelines comprise thin-film multilayered electrodes as described above.

Also, in the transmission line of the present invention, at least one ofthe individual lines of the aforementioned thin lines may be configuredusing a superconductor. In this case, the low-loss characteristics ofthe superconductor advantageously allow a high Q value to be obtained ata level lower than a critical current density.

A resonator according to an embodiment of the present invention isconfigured using the aforementioned transmission line as a resonantline. This allows a resonator having a high unloaded Q value to beobtained.

Also, a filter of an embodiment of the present invention is configuredby providing a signal input/output section in the aforementionedresonator. This provides a filter that is small and produces a smallamount of insertion loss.

In addition, a duplexer of an embodiment of the present invention isconfigured using the aforementioned filter as at least one of atransmitting filter and a receiving filter. By this, a small duplexerthat produces a small amount of insertion loss and that is small can beobtained.

Furthermore, a communication apparatus of an embodiment of the presentinvention is configured using at least one of the aforementioned filtersand duplexer. In such a communication apparatus, insertion losses in REtransmitter and receiver portions can be reduced, and also, quality incommunication with respect to, for example, transmission speeds, can beimproved.

Other features and advantages of the present invention will becomeapparent from the following description of embodiments of the inventionwhich refers to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A and 1B are views showing a configuration of a transmission lineaccording to a first embodiment;

FIGS. 2A and 2B show an example electromagnetic-field distribution inthe transmission line;

FIGS. 3A and 3B show an example electromagnetic-field distribution inanother transmission line;

FIG. 4A shows a portion of the transmission line and FIGS. 4B and 4Cshow exemplary amplitude distributions of current and voltage in thetransmission line;

FIG. 5 shows an analysis model for magnetic-field distributions producedby a line current source;

FIGS. 6A and 6B show magnetic-field strength distributions in theaforementioned model;

FIGS. 7A and 7B show distributions of x components of the magnetic-fieldamplitudes in the aforementioned model;

FIGS. 8A and 8B show strengths of y components of magnetic-fieldamplitudes in positions in the x direction;

FIGS. 9A and 9B show a configuration of a transmission line according toa second embodiment;

FIGS. 10A and 10B show a configuration of a transmission line accordingto a third embodiment;

FIGS. 11A and 11B show a configuration of a transmission line accordingto a fourth embodiment;

FIGS. 12A and 12B show configurations of a transmission line accordingto a fifth embodiment;

FIGS. 13A and 13B show configurations of other transmission linesaccording to the fifth embodiment;

FIGS. 14A and 14B show a configuration of a transmission line accordingto a sixth embodiment;

FIGS. 15A and 15B show a configuration of a transmission line accordingto a seventh embodiment;

FIGS. 16A and 16B show a configuration of a transmission line accordingto an eighth embodiment;

FIG. 17 is an enlarged cross-sectional view of a line portion of atransmission line according to a ninth embodiment;

FIG. 18 is an enlarged cross-sectional view of a line portion of atransmission line according to a tenth embodiment;

FIG. 19 is an enlarged cross-sectional view of a line portion of anothertransmission line according to the tenth embodiment;

FIG. 20 is an enlarged cross-sectional view showing a line portion of atransmission line according to an eleventh embodiment;

FIG. 21A shows a configuration of an exemplary resonator according to atwelfth embodiment;

FIG. 21B shows a configuration of another exemplary resonator accordingto the twelfth embodiment;

FIG. 21C shows a configuration of another exemplary resonator accordingto the twelfth embodiment;

FIG. 21D shows a configuration of another exemplary resonator accordingto the twelfth embodiment;

FIGS. 22A and 22B show a configuration of a filter according to athirteenth embodiment;

FIG. 23 shows a configuration of a duplexer according to a fourteenthembodiment;

FIG. 24 is a block diagram of the aforementioned duplexer; and

FIG. 25 is a block diagram showing a configuration of a communicationapparatus according to a fifteenth embodiment.

DESCRIPTION OF EMBODIMENTS OF THE INVENTION

Hereinbelow, referring to the drawings, a description will be given ofembodiments of a transmission line, a resonator, a filter, a duplexer,and a communication apparatus according to the present invention.

Principles and First Embodiment (FIGS. 1A to 8B)

FIG. 1A is a top view of a configuration of a transmission line, FIG. 1Bis a cross-sectional view along line A—A, and individual enlarged viewsthereof are shown on the right. In FIGS. 1A and 1B, a ground electrode 3(see FIG. 1B) is formed on the entire lower surface of a dielectricsubstrate 1. Thin lines 2 and a continuous line 12 are formed on theupper surface of the dielectric substrate 1. Here, the linewidth of thethin lines 2 is arranged to be substantially the same as the skin depth.

Each of the thin lines 2 is a line having both ends open that has awavelength that is half of the wavelength corresponding to thetransmission frequency of the thin line 2, and they are arranged atidentical pitches in parallel with each other in a state where they areslanted along a signal-propagation direction. The continuous line 12 isformed along the signal-propagation direction so that central portionsof the individual thin lines 2 are connected to each other. In otherwords, the thin lines 2, each having a predetermined length, branch fromboth the left and right sides of the continuous line 12.

FIGS. 2A and 2B show an example of electromagnetic fields and currentdistributions in the thin lines 2. To make the figures clear, however,the figures show a reduced number of the thin lines 2. FIG. 2A showsdistributions of electric fields and magnetic fields in the crosssection through line A—A at a moment when the charge at the left end andthe right end of each thin line 2 is at its maximum. Also, FIG. 2B showsaverage values of z components (in the vertical direction) of themagnetic field that pass through between the individual lines and thecurrent density in the individual lines.

As shown in FIG. 2B, when the individual lines are microscopicallyviewed with current flowing therein, the current density increases atthe individual edge sections. However, when they are viewed in thecross-sectional direction, it is seen that the thin lines 2 are arrangedat a constant pitch and that each thin line 2 has substantially the samelevel of amplitude and phase at both the left and right edges thereof.Therefore, the edge effect is reduced. Specifically, when the thin lines2 are viewed as a single line, the current is distributed effectively inthe form of a sine wave in which the left edge and the right edge arenodes and the central portion is an antinode; and, when viewedmacroscopically, there is effectively no edge effect.

FIGS. 3A and 3B show an example for comparison to FIGS. 2A and 2B andcontaining identical references. In FIGS. 3A and 3B, the linewidth ofeach of the lines shown in FIGS. 2A and 2B is increased to several timesthe skin depth. When the linewidth is thus increased, currentconcentration due to the edge effect on each of the individualconductors becomes apparent, and the loss-reduction effect decreases.

FIG. 4A shows a portion of the transmission line and FIGS. 4B and 4Cshow an example of distributions of voltage and current, respectively,in eight adjacent thin lines. As shown in FIG. 4A, the thin lines 2numbered from 1 to 8 each correspond to a line length of λ_(g)/2 whenthe wavelength is λ_(g). Thus, corresponding to amplitudes of thevoltage and the current carried on the continuous line 12 at the centerof the individual thin lines 2, standing waves of λ_(g)/2 resonance areexcited in the individual thin lines 2.

When the thin lines 2 are viewed as a whole, both the left and rightends of the thin lines 2 that are a half-wavelength in length becomeopen ends. Therefore, they become nodes in the current amplitude, inwhich current is not allowed to flow. Accordingly, there is no currentflowing along the edges of the thin lines 2; thereby the edge effect isreduced. Also, the larger the number of the thin lines 2, the morecontinuous and smooth the current distribution in the thin lines 2;therefore, the edge effect can be efficiently reduced.

Three-dimensional analyses must be performed to obtain the distributionsas shown in FIGS. 2A, 2B, 3A, 3B, 4A, 4B and 4C. FIGS. 4B and 4C showthe distributions for the thin lines numbered from 1 to 8 in FIG. 4A.However, since the calculation volume is very large, rigorous analysesare difficult in practice. Instead, results of static-magnetic-fieldanalyses performed for magnetic-field distributions produced by aplurality of line current sources are shown in which amplitudes andphases are given.

Analysis Model

FIG. 5 shows an analysis model of a plurality of line current sources(i₁, i₂ . . . i_(n)). The model is shown as a cross-sectional view of amulti-microstrip line.

In Model 1 a distribution with the same phase and the same amplitude ofcurrent is used (FIG. 6A), namely:i _(k) =A/√2, (k=1, 2, . . . n)

In Model 2 a distribution with a current phase of 0 to 180° and asinusoidal current amplitude is used (FIG. 6B), namely:i _(k) =A sin{(2k−1)π/2n}, (k=1, 2, . . . n)

Calculation of Magnetic-Field Distribution

Calculation of the magnetic-field distribution is performed according tothe Biot-Savart law.

The magnetic-field vector H produced by a line current source thatpasses through a point (p) on the x-y plane and infinitely continuouslyflowing in a z direction is expressed by the following conventionalformula (1): $\begin{matrix}{H = \frac{\mu_{0}I_{0}e_{z} \times \left( {r - p} \right)}{4{\pi\left( {r - p} \right)}^{2}}} & (1)\end{matrix}$

Accordingly, the magnetic-field distribution produced by a plurality ofline current sources in this model is expressed by the followingconventional formula (2): $\begin{matrix}{H = {\sum\limits_{k}^{\quad}{\frac{\mu_{0}i_{k}}{4\pi}\left( {\frac{e_{z} \times \left( {r - p_{k}} \right)}{\left( {r - p_{k}} \right)^{2}} - \frac{e_{z} \times \left( {r - p_{k}^{(m)}} \right)}{\left( {r - p_{k}^{(m)}} \right)^{2}}} \right)}}} & (2)\end{matrix}$

In the above, p_(k) ^((m)) represents a point of an image position ofp_(k) with the ground electrode as a plane of symmetry. Also, since thecurrent flows in the opposite direction, the second term in the formulahas a negative sign.

Calculation Example

Setting Conditions:

-   Number of lines: n=20-   Total line width: wo=0.5 mm-   Thickness of the substrate: ho=0.5 mm    Coordinates of the line-current source:    x _(k)=[{(2k−1)/2n}−(½)]wo,    and    y _(k) =ho, (where k=1, 2, . . . , n)

FIGS. 6A and 6B show magnetic field strength distributions of Model 1and Model 2, respectively. In the figure, the vertical auxiliary line yrepresents the end of the line group, and the horizontal auxiliary linex represents the boundary surface of the substrate. From a comparison ofthe results, contour lines in the case of Model 2 are not much closer toeach other, the surface current is low, and power loss is smaller in thecase of Model 2.

FIGS. 7A and 7B show distributions of the x component of themagnetic-field amplitude. In the figure, the vertical auxiliary line yrepresents the end of the line group, and the horizontal auxiliary linex represents the boundary surface of the substrate. From a comparison ofthe results, in the case of Model 2, the magnetic-field concentration issmaller, a significant improvement in the edge effect is obtained, andthe loss-reduction characteristics are superior.

Also, FIGS. 8A and 8B show distributions of the y component of themagnetic-field amplitude shown in FIG. 5. In FIGS. 8A and 8B, thevertical auxiliary line y represents the end of the line group, and thehorizontal auxiliary line x represents the boundary surface of thesubstrate. From a comparison of the results, Model 2 is superior inisolation; therefore, it is well suited to integration that is performedin a case where adjacent resonators are provided to configure, forexample, a filter.

Second Embodiment (FIGS. 9A and 9B)

FIGS. 9A and 9B show a plan view and a cross-sectional view,respectively, together with partly-enlarged views thereof, of a secondembodiment. As is apparent from comparison to FIGS. 1A and 1B, in thistransmission line, three continuous lines 12 a, 12 b, and 12 c areformed. The configuration of thin lines 2 is the same as in the case ofFIGS. 1A and 1B. The three continuous lines 12 a, 12 b, and 12 c connectthe thin lines 2 so that corresponding portions along the length of eachof the respective thin lines 2 are connected each other. Thus, identicalportions of each of the individual thin lines 2, which have the samephase, are connected to each other by the three continuous lines 12 a,12 b, and 12 c.

Third Embodiment (FIGS. 10A and 10B)

FIGS. 10A and 10B show a plan view and a cross-sectional view,respectively, together with partly-enlarged views thereof, of a thirdembodiment. As is apparent from comparison to FIGS. 1A and 1B and FIGS.9A and 9B, in this transmission line, three continuous lines 12 a, 12 b,and 12 c are formed; thin lines 2 branch outward only from thecontinuous lines 12 a and 12 c which are arranged at both ends of thegroup of three continuous lines 12 a, 12 b, and 12 c. Also, the centralcontinuous line 12 b is isolated. According to this structure, each ofthe thin lines 2 works as a ¼ wavelength resonant line. The portion ofeach thin line 2 connected to one of the continuous lines 12 a and 12 cfunctions as a short-circuited end, and the other end portion functionsas an open end. The continuous line 12 b functions as a line forpropagating signals.

Fourth Embodiment (FIGS. 11A and 11B)

FIGS. 11A and 11B show a plan view and a cross-sectional view,respectively, together with partly-enlarged views thereof, of a fourthembodiment. As is apparent from comparison to FIGS. 10A and 10B, thethin lines 2 branch symmetrically in bilateral directions from thecontinuous lines 12 a and 12 c on both sides of the group of threecontinuous lines 12 a, 12 b, and 12 c. With this structure, each of thethin lines 2 works as a ¼ wavelength resonant line. The portion of eachthin line 2 connected to one of the continuous lines 12 a and 12 cfunctions as a short-circuited end, and the other end portion functionsas an open end. The continuous line 12 b functions as a line forpropagating signals.

Fifth Embodiment (FIGS. 12A to 13B)

FIGS. 12A, 12B, 13A and 13B show plan views of four transmission linesthat have four different patterns of thin lines 2. FIGS. 12A and 12Bshow two examples where the thin lines 2 branch diagonally in an upperright direction and a lower left direction from the continuous line 12;and FIGS. 13A and 13B show two examples in which the thin lines 2 branchsymmetrically with respect to the central continuous line 12 as thesymmetry axis.

In FIGS. 12A and 13A, the thin lines 2 are curved concavely toward theline 12. In FIGS. 12B and 13B, the thin lines 2 are curved convexlytoward the line 12.

In any one of the transmission lines, by forming the thin lines 2 to becurved, capacitive coupling and mutual dielectric coupling between thethin lines 2 can be controlled more freely than in the case where thethin lines 2 are formed linearly. Also, this allows electrical lengthsof the thin lines 2 to be adjusted while keeping their overall physicallength constant.

Sixth Embodiment (FIGS. 14A and 14B)

FIGS. 14A and 14B show a plan view and a cross-sectional view,respectively, together with partly-enlarged views thereof, of a sixthembodiment. As seen in FIG. 14B, unlike the lines shown in FIGS. 1A and1B, thin lines 2 and a continuous line 12 are individually configured ofslotted lines. In a transmission line composed of these slotted lines,the current concentration at the end is also reduced, and thetransmission loss is reduced.

Seventh Embodiment (FIGS. 15A and 15B)

FIGS. 15A and 15B show a plan view and a cross-sectional view,respectively, together with partly-enlarged views thereof, of a seventhembodiment. In this example, two continuous lines 12 a and 12 b areprovided. These continuous lines 12 a and 12 b are provided so that eachend of each of the thin lines 2 is connected to a respective one of thelines 12 a and 12 b. According to this structure, each of the thin lines2 works as a half-wavelength line in which both ends areshort-circuited, and the continuous lines 12 a and 12 b connect antinodeportions of the current amplitudes.

Eighth Embodiment (FIGS. 16A and 16B)

FIGS. 16A and 16B show a plan view and a cross-sectional view,respectively, together with partly-enlarged views thereof, of an eighthembodiment. In this example, as seen in FIG. 16B, two continuous lines12 a and 12 b and thin lines 2 are individually configured of slottedlines, and these continuous lines 12 a and 12 b are provided so thateach end of each of the thin lines 2 is connected to a respective one ofthe lines 12 a and 12 b. In a transmission line composed of theseslotted lines, the current concentration at the end is also reduced, andthe transmission loss is reduced.

Ninth Embodiment (FIG. 17)

FIG. 17 is an enlarged view of line portions of a ninth embodiment. Theline width of each line is substantially the same as or smaller than theskin depth of the conductor. According to this, current flows formaintaining a magnetic flux that passes through a gap (space) betweenthe right side of one conductor and the left side of the next conductor.This spacing is such that the left side current and the right sidecurrent interfere with each other. By this, reactive current that has aphase deviated from a resonant phase can be reduced; and as a result,the power loss can be significantly reduced.

Tenth Embodiment (FIGS. 18 and 19)

FIG. 18 is an enlarged view of line portions of a tenth embodiment. Inthis example, a thin-film conductor layer, a thin-film dielectric layer,a thin-film conductor layer, and a thin-film dielectric layer areoverlaid on a surface of a dielectric substrate in that order. Inaddition, a conductor layer is provided as the top layer; thusconfiguring the line as a thin-film multilayered electrode in athree-layer structure. In this way, since the thin lines 2 aremultilayered in the film-thickness direction, the skin effect at thesurface of the substrate can be reduced, and the conductor loss can befurther reduced.

FIG. 19 shows a case where a dielectric material is filled in each gapbetween the above-described thin-film multilayered electrodes. Accordingto this structure, short-circuiting between the adjacent lines andshort-circuiting between the layers can be easily prevented, therebyallowing improvement in reliability and stabilization in characteristicsto be implemented.

Eleventh Embodiment (FIG. 20)

FIG. 20 is an enlarged view of conductor portions of an eleventhembodiment. In this example, superconductors are used as electrodes. Forexample, an Yttrium-group or Bismuth-group high-temperaturesuperconductor material is used. Generally, when a superconductormaterial is used for the electrodes, an upper limit of the currentdensity must be determined so that withstand-power characteristics arenot reduced. However, according to the configuration with one continuousline and the plurality of thin lines branched therefrom, the lineportions have no substantial edge sections. Therefore, no significantcurrent concentration occurs, allowing operation to be easily performedat a level lower than the critical current density of thesuperconductor. As a result, low-loss characteristics of thesuperconductor can be efficiently used.

Twelfth Embodiment (FIGS. 21A to 21D)

FIGS. 21A to 21D show four examples of resonators that use theabove-described transmission lines as resonant lines. In FIGS. 21A and21C, there are shown examples in each of which thin lines 2 are formedin bilateral symmetry with respect to a central continuous line 12. InFIGS. 21B and 21D, each structure is such that central portions oflinear thin lines 2 are connected to each other via a continuous line12. In the examples shown in FIGS. 21A and 21B, end terminals 13 and 14are formed such that line lengths of the thin lines 2 are all the same.In the examples shown in FIGS. 21C and 21D, end terminals 13 and 14 areformed only at both ends of the resonant line.

Thirteenth Embodiment (FIGS. 22A and 22B)

FIGS. 22A and 22B show a configuration of a filter, in which FIG. 22Ashows a top view of a dielectric substrate 1 on which resonant lines areformed, and FIG. 22B is a side view of the entire configuration of thefilter. On an upper surface of the dielectric substrate 1, there arcarranged four transmission lines that are similar to those shown in FIG.21D; and external-coupling electrodes 5 for capacitively coupling therespective resonant lines are formed at both ends. The external-couplingelectrodes 5 extend to a front surface (outer surface) as an inputterminal and an output terminal. Ground electrodes are formed on a lowersurface and four peripheral surfaces of the dielectric substrate 1.Also, another dielectric substrate 1′ having ground electrodes formed onan upper surface and four peripheral surfaces is formed on thedielectric substrate 1. By this, a filter using triplet-structuredresonators is configured.

According to the above-described structure, adjacent resonators aredielectrically coupled, thereby, a filter that is formed of fourresonators and that provides bandpass characteristics is obtained.

Fourteenth Embodiment (FIGS. 23 and 24)

FIG. 23 is a view showing the configuration of a duplexer and is a topview showing a state where an upper shield cover is removed. In thefigure, 10 and 11 denote filters each having the configuration of thedielectric substrate portion shown in FIGS. 22A and 22B. The filter 10is used as a transmitting filter, and the filter 11 is used as areceiving filter. The filters 10 and 11 are mounted on an upper surfaceof an insulating substrate 6. On the substrate 6, there are formed abranching line 7, an ANT terminal, a TX terminal, and an RX terminal, towhich external-coupling electrodes of the filters 10 and 11 andelectrode portions of the substrate 6 are wire-bonded. A groundelectrode GND is formed substantially on the entire lower surfaceexcluding the terminal portions of the insulating substrate 6. Theshield cover is mounted on the upper portion indicated by dotted linesin the figure.

FIG. 24 is a block diagram of the duplexer. According to this structure,intrusion of transmitted signals to a receiver circuit and intrusion ofreceived signals to a transmitter circuit can be prevented. Also,transmitted signals from the transmitter circuit are limited by atransmitting filter to a transmitting-frequency band and are guided toan antenna; and received signals from the antenna are limited by areceiving filter to a receiving-frequency band and are fed to areceiver.

Fifteenth Embodiment (FIG. 25)

FIG. 25 is a block diagram of the configuration of a communicationapparatus according to a fifteenth embodiment. In this, a duplexer hasthe configuration shown in FIGS. 23 and 24, comprising a transmittingfilter and a receiving filter. A transmitter circuit and a receivercircuit are configured on a circuit substrate, the transmitter circuitis connected to the TX terminal, the receiver circuit is connected tothe RX terminal, and an antenna is connected to the ANT terminal. Inthis way, the duplexer is mounted on the aforementioned circuitsubstrate.

Although embodiments of the transmission line, resonator, filter,duplexer, and communication apparatus according to the present inventionhave been described, it is to be understood that the invention is notrestricted to the described embodiments. On the contrary, the inventionis intended to cover various other modifications and equivalentarrangements included within the spirit and scope of the invention.

1. A transmission line comprising a continuous line and a plurality ofthin lines each branching from said continuous line each having apredetermined common overall length, wherein the branching direction ofeach of said plurality of thin lines is slanted with respect to saidcontinuous line.
 2. The transmission line according to claim 1, whereinthe thin lines on each side of the continuous line are substantiallyparallel to each other.
 3. A transmission line comprising: a continuousline and a plurality of thin lines each branching from said continuousline and each havinR a predetermined common overall length; and firstand second connecting lines each on a respective side of andsubstantially parallel to said continuous line, wherein said pluralityof thin lines on each side of the continuous line are connected to eachother by the corresponding one of said connecting lines, and saidcorresponding connecting line connects corresponding portions of each ofsaid plurality of thin lines on said respective side.
 4. A transmissionline comprising a continuous line and a plurality of thin lines eachbranching from said continuous line and each having a predeterminedcommon overall length, wherein said plurality of thin lines are curvedlines, and either capacitive coupling or mutual dielectric couplingexists between each adjacent pair of said thin lines.
 5. A transmissionline comprising a continuous line and a plurality of thin lines eachbranching from said continuous line and each having a predeterminedcommon overall length, wherein each of said plurality of thin linescomprises a thin-film multilayered electrode formed by overlayingthin-film dielectric layers and thin-film conductor layers.
 6. Thetransmission line according to claim 5, further comprising a dielectricmaterial filled in a gap between each pair of said plurality of thinlines that are adjacent to each other.
 7. A resonator comprising: atransmission line comprising a continuous line and a plurality of thinlines each branching from said continuous line and each having apredetermined common overall length; and a pair of terminals connectedrespectively to ends of said continuous line.
 8. A filter comprising aplurality of resonators according to claim 7, an input terminal coupledto a terminal of one of said resonators; and an output terminal coupledto a terminal of another one of said resonators.
 9. A duplexercomprising a transmitting filter and a receiving filter, an outputterminal of said transmitting filter and an input terminal of saidreceiving filter being connected in common to an antenna terminal, atleast one of said transmitting and receiving filters being a filteraccording to claim
 8. 10. A communication apparatus comprising theduplexer according to claim 9; a transmitting circuit connected to aninput terminal of said transmitting filter; and a receiving circuitconnected to an output terminal of said receiving filter.
 11. Acommunication apparatus comprising: a high frequency circuit comprisingat least one of a transmitting circuit and a receiving circuit; and thefilter according to claim 8, wherein at least one of said input terminaland said output terminal of said filter is connected to said highfrequency circuit.